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参数资料
| 型号： | AD8314 |
| 厂商： | Analog Devices, Inc. |
| 英文描述： | 100 MHz-2500 MHz 45 dB RF Detector/Controller |
| 中文描述： | 100兆赫，2500兆赫45分贝射频检测器/控制器 |
| 文件页数： | 11/16页 |
| 文件大小： | 274K |
| 代理商： | AD8314 |
AD8314
–11–
REV. 0
The relationship between the input level and the setpoint voltage
follows from the nominal transfer function of the device (V
UP
vs.
Input Amplitude, see Figure 1). For example, a voltage of 1 V
on VSET is demanding a power level of 0 dBm at RFIN. The cor-
responding power level at the output of the power amplifier will be
greater than this amount due to the attenuation through the direc-
tional coupler.
When connected in a PA control loop, as shown in Figure 31,
the voltage V
UP
is not explicitly used, but is implicated in again
setting up the required averaging time, by choice of C
F
. However,
now the effective loop response time is a much more complicated
function of the PA’s gain-control characteristics, which are very
nonlinear. A complete solution requires specific knowledge of
the power amplifier.
The transient response of this control loop is determined by the
filter capacitor, C
F
. When this is large, the loop will be uncon-
ditionally stable (by virtue of the “dominant pole” generated
by this capacitor), but the response will be sluggish. The minimum
value ensuring stability should be used, requiring full attention
to the particulars of the power amplifier control function. Because
this is invariably nonlinear, the choice must be made for the
worst-case condition, which usually corresponds to the smallest
output from the PA, where the gain function is steepest. In practice,
an improvement in loop dynamics can often be achieved by adding
a response zero, formed by a resistor in series with C
F
.
Power-On and Enable Glitch
As already mentioned, the AD8314 can be put into a low power
mode by pulling the ENBL pin to ground. This reduces the quies-
cent current from 4.5 mA to 20
μ
A. Alternatively, the supply can
be turned off completely to eliminate the quiescent current. Figures
13 and 23 show the behavior of the V_DN output under these
two conditions (in Figure 23, ENBL is tied to VPOS). The glitch
that results in both cases can be reduced by loading the V_DN
output.
Input Coupling Options
The internal 5 pF coupling capacitor of the AD8314, along with
the low frequency input impedance of 2 k
give a high-pass input
corner frequency of approximately 16 MHz. This sets the mini-
mum operating frequency. Figure 32 shows three options for
input coupling. A broadband resistive match can be implemented
by connecting a shunt resistor to ground at RFIN (Figure 32a).
This 52.3
resistor (other values can also be used to select dif-
ferent overall input impedances) resistor combines with the
input impedance of the AD8314 (3 k
i
2 pF) to give a broad-
band input impedance of 50
. While the input resistance and
capacitance (C
IN
and R
IN
) will vary by approximately
±
20% from
device to device, the dominance of the external shunt resistor
means that the variation in the overall input impedance will
be close to the tolerance of the external resistor.
At frequencies above 2 GHz, the input impedance drops below
250
(see Figure 9), so it is appropriate to use a larger value of
shunt resistor. This value is calculated by plotting the input
impedance (resistance and capacitance) on a Smith Chart and
choosing the best value of shunt resistor to bring the input imped-
ance closest to the center of the chart. At 2.5 GHz, a shunt
resistor of 165
is recommended.
A reactive match can also be implemented as shown in Figure
32b. This is not recommended at low frequencies as device toler-
ances will dramatically vary the quality of the match because of
the large input resistance. For low frequencies, Option a or
Option c (see below) are recommended.
In Figure 32b, the matching components are drawn as general
reactances. Depending on the frequency, the input impedance at
that frequency and the availability of standard value components,
either a capacitor or an inductor will be used. As in the previous
case, the input impedance at a particular frequency is plotted on
a Smith Chart and matching components are chosen (shunt
or series L, shunt or series C) to move the impedance to the
center of the chart. Table II gives standard component values
for some popular frequencies. Matching components for other
frequencies can be calculated using the input resistance and
reactance data over frequency which is given in Figure 9. Note
that the reactance is plotted as though it appears in parallel with
the input impedance (which it does because the reactance is prima-
rily due to input capacitance).
The impedance matching characteristics of a reactive matching
network provide voltage gain ahead of the AD8314; this increases
the device sensitivity (see Table II). The voltage gain is calculated
using the equation:
Voltage Gain
R
R
dB
=
20
2
1
10
log
where
R
2 is the input impedance of the AD8314 and
R
1 is the
source impedance to which the AD8314 is being matched. Note
that this gain will only be achieved for a perfect match. Component
tolerances and the use of standard values will tend to reduce
the gain.
R
52.3
V
C
IN
AD8314
50
V
50
V
SOURCE
R
IN
C
C
RFIN
V
BIAS
a. Broadband Resistive
50
V
SOURCE
C
IN
AD8314
50
V
R
IN
C
C
RFIN
V
BIAS
X2
X1
b. Narrowband Reactive
C
IN
AD8314
R
IN
C
C
RFIN
V
BIAS
R
ATTN
STRIPLINE
c. Series Attenuation
Figure 32. Input Coupling Options
Figure 32c shows a third method for coupling the input signal
into the AD8314, applicable in applications where the input signal
is larger than the input range of the log amp. A series resistor,
connected to the RF source, combines with the input impedance
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