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参数资料
| 型号： | AD8316ACP-EVAL |
| 厂商： | Analog Devices, Inc. |
| 英文描述： | Dual Output GSM PA Controller |
| 中文描述： | 双输出GSM功率放大器控制器 |
| 文件页数： | 13/20页 |
| 文件大小： | 497K |
| 代理商： | AD8316ACP-EVAL |
REV. C
AD8316
–13–
AD8316
RFIN
OUT2
FLT2
VPOS
OUT1
COMM
FLT1
BSEL
ENBL
VSET
1
2
3
4
5
10
9
8
7
6
RFIN
+V
S
V
SET
C
FLT1
V
BSEL
V
OUT1
V
OUT2
C
FLT2
R1
52.3
C1
0.1 F
+V
2.7 TO 5.5V
Figure 6. Basic Connections (Shown with MSOP Pinout)
OUT1
DIRECTIONAL
COUPLER
OUT2
FLT2
VSET
BSEL
FLT1
RFIN
R1
52.3
C
FLT1
C
FLT2
ATTN
DAC
GAIN
CONTROL
VOLTAGES
RFIN1
BAND
SELECT
RFIN2
RX1
RX2
TX1
TX2
ANT
PWR
AMP
Figure 7. Block Diagram of Typical Application
A supply voltage of 2.7 V to 5.5 V is required for the AD8316.
The supply to the VPOS pin should be decoupled with a low
inductance 0.1
μ
F surface-mount ceramic capacitor close to the
device. The AD8316 has an internal input coupling capacitor,
which negates the need for external ac coupling. This capacitor,
along with the device’s low frequency input impedance of approxi-
mately 3.0 k
, sets the minimum usable input frequency to around
20 MHz. A broadband 50
input match is achieved in this
example by connecting a 52.3
resistor between RFIN and
ground (COMM). A plot of input impedance versus frequency
is shown TPC 9. Other matching methods are also possible
(see the Input Coupling Options section).
In a power control loop, the AD8316 provides both the detector
and controller functions.
A number of options exist for coupling the RF signal from the
power amplifiers (PA) to the AD8316 input. Because only one
PA output is active at any time, a single RF input on the
AD8316 is sufficient in all cases.
Two directional couplers can be used directly at the PA outputs.
The outputs of these couplers would be passively combined
before being applied to the AD8316 RF input (in general,
some additional attenuation will be required between the coupler
and the AD8316). Another option involves using a dual-direc-
tional coupler between the PA and T/R switch. This device has
two inputs/outputs and a single-coupled output so that no exter-
nal combiner is required.
A third option is to use a single broadband directional coupler
at the output of the transmit/receive (T/R) switch (the outputs
from the two PAs are combined in the T/R switch). This is
shown in Figure 7. This provides the advantage of enabling the
power at the output of the T/R switch to be precisely set, elimi-
nating any errors due to insertion loss and insertion loss
variations of the T/R switch.
A setpoint voltage is applied to VSET from the controlling
source, generally a DAC. Any imbalance between the RF input
level and the level corresponding to the setpoint voltage will be
corrected by the selected output, OUT1 or OUT2, which drives
the gain control terminal of the PAs. This restores a balance
between the actual power level sensed at the input of the AD8316
and the demanded value determined by the setpoint. This assumes
that the gain control sense of the variable gain element is posi-
tive; that is, an increasing voltage from OUT1 or OUT2 will
tend to increase gain. The outputs can swing from 100 mV
above ground to within 100 mV of the supply rail and can source
up to 12 mA. (A plot of maximum output voltage versus output
current is shown in TPC 19.) OUT1/OUT2 are capable of
sinking more than 200
μ
A.
Range on VSET and RF Input
The relationship between RF input level and the setpoint volt-
age follows from the nominal transfer function of the device (see
TPCs 2, 3, 5, and 6). At 0.9 GHz, for example, a voltage of 1 V
on VSET indicates a demand for –17 dBm (–30 dBV) at RFIN.
The corresponding power level at the output of the power ampli-
fier will be greater than this amount due to the attenuation
through the directional coupler. For setpoint voltages of less
than approximately 200 mV and RF input amplitudes greater
than approximately –50 dBm, V
OUT
will remain unconditionally
at its minimum level of approximately 250 mV. This feature can
be used to prevent any spurious emissions during power-up and
power-down phases. Above 250 mV, VSET will have a linear
control range up to 1.4 V, corresponding to a dynamic range of
49 dB. This results in a slope of 22.2 mV/dB or approximately
45.5 dB/V.
Transient Response
The time domain response of power amplifier control loops,
using any kind of controller, is only partially determined by the
choice of filter which, in the case of the AD8316, has a true
integrator form 1/sT, as shown in Equation 7, with a time con-
stant given by Equation 8. The large signal step response is also
strongly dependent on the form of the gain control law. Never-
theless, some simple rules can be applied. When the filter capacitor
C
FLT
is very large, it will dominate the time domain response,
but the incremental bandwidth of this loop will still vary as
V
OUT
traverses the nonlinear gain control function of the PA, as
shown in Figure 5. This bandwidth will be highest at the
point where the slope of the tangent drawn on this curve is
greatest—that is, for power outputs near the center of the PA’s
range—and will be much reduced at both the minimum and the
maximum power levels, where the slope of the gain control
curve is lowest, due to its S-shaped form. Using smaller values
of C
FLT
, the loop bandwidth will generally increase, in inverse
proportion to its value. Eventually, however, a secondary effect
will appear, due to the inherent phase lag in the power amplifier’s
control path, some of which may be due to parasitic or deliber-
ately added capacitance at the OUT1 and OUT2 pins. This results
in the characteristic poles in the ac loop equation moving off the
real axis and thus becoming complex (and somewhat resonant).
This is a classic aspect of control loop design.
The lowest permissible value of C
FLT
needs to be determined
experimentally for a particular amplifier and circuit board lay-
out. For GSM and DCS power amplifiers, C
FLT
will typically
range from 150 pF to 300 pF.
In many cases, some improvement in the worst-case response
time can be achieved by including a small resistance in series with
C
FLT
; this generates an additional zero in the closed-loop trans-
fer function, which will serve to cancel some of the higher-order
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