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| 型号： | LT1766HGN#TR |
| 厂商： | LINEAR TECHNOLOGY CORP |
| 元件分类： | 稳压器 |
| 英文描述： | 3 A SWITCHING REGULATOR, 228 kHz SWITCHING FREQ-MAX, PDSO16 |
| 封装： | 0.150 INCH, PLASTIC, SSOP-16 |
| 文件页数： | 5/28页 |
| 文件大小： | 284K |
| 代理商： | LT1766HGN#TR |
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13
LT1766/LT1766-5
1766fb
Discontinuous mode is entered when the output load
current is less than one-half of the inductor ripple current
(ILP-P). In this mode, inductor current falls to zero before
the next switch turn on (see Figure 8). Buck converters will
be in discontinuous mode for output load current given by:
IOUT
Discontinuous Mode
The inductor value in a buck converter is usually chosen
large enough to keep inductor ripple current (ILP-P) low;
this is done to minimize output ripple voltage and maxi-
mize output load current. In the case of large inductor
values, as seen in the equation above, discontinuous
mode will be associated with “light loads.”
When choosing small inductor values, however, discon-
tinuous mode will occur at much higher output load
currents. The limit to the smallest inductor value that can
be chosen is set by the LT1766 peak switch current (IP)
and the maximum output load current required, given by:
IOUT(MAX)
Discontinuous Mode
Example: For VIN = 15V, VOUT = 5V, VF = 0.63V, f = 200kHz
and L = 10
H.
IOUT(MAX)
Discontinuous
Mode
IOUT(MAX)
= 0.639A
Discontinuous Mode
What has been shown here is that if high inductor ripple
current and discontinuous mode operation can be toler-
ated, small inductor values can be used. If a higher output
load current is required, the inductor value must be
increased. If IOUT(MAX) no longer meets the discontinuous
mode criteria, use the IOUT(MAX) equation for continuous
mode; the LT1766 is designed to operate well in both
modes of operation, allowing a large range of inductor
values to be used.
APPLICATIO S I FOR ATIO
WU
UU
Short-Circuit Considerations
The LT1766 is a current mode controller. It uses the VC
node voltage as an input to a current comparator which
turns off the output switch on a cycle-by-cycle basis as
this peak current is reached. The internal clamp on the VC
node, nominally 2V, then acts as an output switch peak
current limit. This action becomes the switch current limit
specification. The maximum available output power is
then determined by the switch current limit.
A potential controllability problem could occur under
short-circuit conditions. If the power supply output is
short circuited, the feedback amplifier responds to the low
output voltage by raising the control voltage, VC, to its
peak current limit value. Ideally, the output switch would
be turned on, and then turned off as its current exceeded
the value indicated by VC. However, there is finite response
time involved in both the current comparator and turnoff
of the output switch. These result in a minimum on time
tON(MIN). When combined with the large ratio of VIN to
(VF + I R), the diode forward voltage plus inductor I R
voltage drop, the potential exists for a loss of control.
Expressed mathematically the requirement to maintain
control is:
ft
VI R
V
ON
F
IN
≤
+
where:
f = switching frequency
tON = switch minimum on time
VF = diode forward voltage
VIN = Input voltage
I R = inductor I R voltage drop
If this condition is not observed, the current will not be
limited at IPK, but will cycle-by-cycle ratchet up to some
higher value. Using the nominal LT1766 clock frequency
of 200KHz, a VIN of 40V and a (VF + I R) of say 0.7V, the
maximum tON to maintain control would be approximately
90ns, an unacceptably short time.
The solution to this dilemma is to slow down the oscillator
when the FB pin voltage is abnormally low thereby indicat-
ing some sort of short-circuit condition. Oscillator fre-
quency is unaffected until FB voltage drops to about 2/3 of
<
+
()(
–
)
()(
)( )()
VV
V
Vf L
OUT
F
IN
OUT
F
IN
2
=
() (
)
+
I
If L V
VV
V
P
PIN
OUT
F
IN
OUT
F
2
()(
)
()( )(
)
()(
–
)
ILP-P
=
+
( . ) ()(
)( )
(.
)(
– – .
)
–
1 5
200 10
10
15
2 5 0 63 1550 63
23
5
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