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参数资料
| 型号： | 74017 |
| 厂商： | Skyworks Solutions Inc. |
| 英文描述： | On the Direct Conversion Receiver |
| 中文描述： | 在直接转换接收器 |
| 文件页数： | 8/10页 |
| 文件大小： | 315K |
| 代理商： | 74017 |
On the Direct Conversion Receiver
CX74017
101735A
Skyworks Solutions, Inc., Proprietary and Confidential
7
July 20, 2001
Preliminary Data Subject to Change
It can be seen from these equations and in Figure 13 that the
DC component due to the second order non-linearity is growing
with twice the slope of the fundamental on a logarithmic scale.
At the intercept point,
IIP2
a
A
a
A
a
=
=
2
1
2
Due to the doubled slope of the second-order product,
IM2
Pout
with
Pin
IIP2
=
+
=
Noise
Low frequency noise becomes a great concern in DCR [14], as
significant gain is allocated to baseband stages after the mixer.
Weak baseband signal levels of a few millivolts are still very
vulnerable to noise. This requires stronger RF stage gain to
alleviate the poor noise figure of baseband blocks, but of course
this must be traded against the linearity problems, just
described, that accompany higher RF gain.
Flicker noise, that is, 1/f noise, is the major baseband noise
contributor. Associated with DC flow, it has a spectral response
proportional to 1/f. In RF circuits, 1/f noise tends to be
modulated onto the RF signal. In the case of a mixer with
baseband output, 1/f noise sees especially high conversion gain.
In practice, flicker noise becomes an issue for Metal Oxide
Semiconductor (MOS) devices more than bipolar, and is
modeled as a voltage source in series with the gate. 1/f noise
complicates the use of MOS transistors for RF circuits, since the
main method of reducing it in MOS is to increase the transistor’s
size, which increases the device capacitance. This adversely
affects RF gain. For this reason, it is preferable to use bipolar
transistors for DCR mixer designs. In the first baseband stages
after the mixer, it becomes possible to use MOS devices, as the
transistor-size tradeoff is feasible at low frequencies.
I/Q mismatches
Due to the high frequency of the LO, it is not possible to
implement the IQ demodulator digitally. An analog IQ
demodulator exhibits gain and phase imbalances between the
two branches, as well as the introduction of DC offsets. Such
imperfections distort the recovered constellation. Assuming
α, being the amplitude and phase mismatch respectively
between the demodulator quadrature ports, and the complex
signal incident upon it having in-phase and quadrature
components I and Q:
()
() (
)
sin(
1
2
)
sin(
)
cos(
)
cos(
2
)
sin(
)
cos(
ω
α
ω
+
+
=
+
=
t
Q
t
I
Qout
t
Q
t
I
Iout
Filtering out the high frequency terms:
()(
))
cos(
)
sin(
1
α
Q
I
Qout
I
Iout
+
+
=
101735A 13_071801
1
2
1
Pin
Pout
fu
nd
am
en
tal
D
C
(2
nd
or
de
r)
IP2
IIP2
OIP2
IP3
3
rd
or
de
r
1
3
1
2
1
Pin
Pout
fu
nd
am
en
tal
D
C
(2
nd
or
de
r)
IP2
IIP2
+49dBm
OIP2
Pin
Pout
Figure 13. Second Order Intercept Point (IP2)
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