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参数资料
| 型号： | lCL8052CDD |
| 厂商： | Intersil Corporation |
| 英文描述： | Precision 4 1/2 Digit, A/D Converter |
| 中文描述： | 精密4 1 / 2位，A / D转换 |
| 文件页数： | 11/19页 |
| 文件大小： | 168K |
| 代理商： | LCL8052CDD |
3-44
ICL8052 vs ICL8068
The ICL8052 offers significantly lower input leakage currents
than the ICL8068, and may be found preferable in systems
with high input impedances. However, the ICL8068 has
substantially lower noise voltage, and is the device of choice
for systems where noise is a limiting factor, particularly in low
signal level conditions.
Max Clock Frequency
The maximum conversion rate of most dual-slope A/D
converters is limited by frequency response of the compara-
tor. The comparator in this circuit is no exception, even
though it is entirely NPN with an open-loop, gain-bandwidth
product of 300MHz. The comparator output follows the inte-
grator ramp with a 3
μ
s delay, and at a clock frequency of
160kHz (6
μ
s period) half of the first reference integrate clock
period is lost in delay. This means that the meter reading will
change from 0 to 1 with 50
μ
V input, 1 to 2 with 150
μ
V, 2 to 3
at 250
μ
V, etc. This transition at midpoint is considered
desirable by most users. However, if the clock frequency is
increased appreciably above 160kHz, the instrument will
flash “1” on noise peaks even when the input is shorted.
For many dedicated applications where the input signal is
always on one polarity, the dealy of the comparator need not
be limitation. Since the non-linearity and noise do not
increase substantially with frequency, clock rates of up to
approximately 1MHz may be used. For a fixed clock
frequency, the extra count or counts caused by comparator
delay will be a constant and can be subtracted out digitally.
The minimum clock frequency is established by leakage on
the auto-zero and reference caps. With most devices,
measurement cycles as long as 10 seconds give no measur-
able leakage error.
To achieve maximum rejection of 60Hz pickup, the signal
integrate cycle should be a multiple of 60Hz. Oscillator
frequencies of 300kHz, 200kHz, 150kHz, 120kHz, 100kHz,
40kHz, 33
1
/
3
kHz, etc, should be selected. For 50Hz
rejection, oscillator frequencies of 250kHz, 166
2
/
3
kHz,
125kHz, 100kHz, etc. would be suitable. Note that 100kHz
(2.5 readings/second) will reject both 50Hz and 60Hz.
The clock used should be free from significant phase or
frequency jitter. A simple two-gate oscillator and one based
on CMOS 7555 timer are shown in the Applications section.
The multiplexed output means that if the display takes
significant current from the logic supply, the clock should
have good PSRR.
Applications
Specific Circuits Using the 8068/71C03 8052/71C03
Figure 7 shows the complete circuit for a
±
4
1
/
2
digit
(
±
200mV full scale) A/D converter with LED readout using
the internal reference of the 8068/52. If an external
reference is used, the reference supply (pin 7) should be
connected to ground and the 300pF reference cap deleted.
The circuit also shows a typical RC input filter. Depending on
the application, the time-constant of this filter can be made
faster, slower, or the filter deleted completely. The
1
/
2
digit
LED is driven from the 7-segment decoder, with a zero
reading blanked by connecting a D
5
signal to RBI input of
the decoder.
A voltage translation network is connected between the com-
parator output of the 8068/52 and the auto-zero input of the
71C03. The purpose of this network is to assure that, during
auto-zero, the output of the comparator is at or near the
threshold of the 71C03 logic (+2.5V) while the auto-zero
capacitor is being charged to V
REF
(+100mV for a 200mV
instrument). Otherwise, even with 0V in, some reference inte-
grate period would be required to drive the comparator output
to the threshold level. This would show up as an equivalent
offset error. Once the divider network has been selected, the
unit-to-unit variation should contribute less than a tenth of a
count error. A second feature is the back-to-back diodes, used
to lower the noise. In the normal operating mode they offer a
high impedance and long integrating time constant to any
noise pulses charging the auto-zero cap. At startup or recov-
ery from an overload, their impedance is low to large signals
so that the cap can be charged up in one auto-zero cycle. The
buffer gain does not have to be set precisely at 10 since the
gain is used in both the integrate and deintegrate phase. For
scale factors other then 200mV the gain of the buffer should
be changed to give a
±
2V buffer output. For 2.0000V full scale
this means unity gain and for 20,000mV (1
μ
V resolution) a
gain of 100 is necessary. Not all 8068As can operate properly
at a gain of 100 since their offset should be less than 10mV in
order to accommodate the auto-zero circuitry. However, for
devices selected with less than 10mV offset, the noise perfor-
mance is reasonable with approximately 1.5
μ
V near full scale.
On all scales less than 200mV, the voltage translation network
should be made adjustable as an offset trim.
The auto-zero cap should be 1
μ
F for all scales and the refer-
ence capacitor should be 1
μ
F times the gain of the buffer
amplifier. At this value if the input leakages of the 8052/8068
are equal, the droop effects will cancel giving zero offset.
This is especially important at high temperature. Some
typical component values are shown in Table 1. For 3
1
/
2
digit
conversion, use 12kHz clock.
V++ = +15V, V+ = 5V, V- = -15V
Clock Freq. = 120kHz (4
1
/
2
Digit) or 12kHz (3
1
/
2
Digit)
TABLE 1.
SPECIFICATION
VALVE
UNITS
Full Scale V
IN
20
200
2000
mV
Buffer Gain
100
(See
Note)
10
1
V/V
R
INT
100
100
100
k
C
INT
0.22
0.22
0.22
μ
F
C
AZ
1.0
1.0
1.0
μ
F
C
REF
10
10
1.0
μ
F
V
REF
Resolution (4
1
/
2
Digit)
10
100
1000
mV
1
10
100
μ
V
NOTE: Comment on offset limitations above. Buffer gain does not
improve ICL8052 noise performance adequately.
)
---------------+
ICL8052/ICL71C03, ICL8068/ICL71C03
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